System and method for code division multiplexed optical communication

ABSTRACT

A system for optical communication forms a family of orthogonal optical codes modulated by a data stream. The orthogonal codes are formed by creating a stream of evenly spaced-apart pulses using a pulse spreader circuit and modulating the pulses in amplitude and/or phase to form a family of orthogonal optical code words, each representing a symbol. A spreader calibration circuit is used to ensure accurate timing and modulation. Each code word is further modulated by a predetermined number of data bits. The data modulation scheme splits a code word into H and V components, and further processes the components prior to modulation with data, followed by recombining with a polarization beam combiner. The data-modulated code word is then sent, along with others to receiver. The received signal is detected and demodulated with the help of a symbol synchronization unit which establishes the beginning and end of the code words. A polarization mode distortion compensator at the receiver cooperates with a state of polarization compensator at the transmitter to mitigate polarization distortion in the fiber.

RELATED APPLICATIONS

The present application claims priority to U.S. Provisional applicationNo. 60/234,930, filed Sep. 26, 2000.

FIELD OF THE INVENTION

The present invention relates to the field of optical communicationsystems utilizing modulation techniques to obtain high spectralefficiency.

BACKGROUND OF THE INVENTION

Dense wavelength division multiplexing (DWDM) increases the capacity ofembedded fiber by assigning incoming optical signals to specificfrequencies (wavelength, lambda) within a designated frequency band andthen multiplexing the resulting signals out onto one fiber. DWDMcombines multiple optical signals so that they can be amplified as agroup and transported over a single fiber to increase capacity of thetelecommunication network. Each signal carried can be at a differentrate (OC-3112/24, etc.) and in a different format (SONET, ATM, data,etc.). Limiting bandwidth of the useable band of the optical fiber toaccommodate future growth is the driving force behind the effort toincrease the spectral efficiency of DWDM systems.

FIG. 1 is a block diagram of a prior art simplex DWDM system. A DWDMmultiplexer 110 combines several optical signals, hereinafter referredto as channels, into a single multi-channel optical signal that istransmitted through the optical fiber 120. Optical amplifiers 125 may beconnected to the optical fiber 120 to amplify the optical signal.Conversely, the DWDM demultiplexer 130 receives the multi-channeloptical signal transmitted through the optical fiber 120 and splits itinto separate channels. Each channel is characterized by a distinctwavelength designated as λ_(i) in FIG. 1 where the index, i, runs from 1to N where N is the number of channels in the DWDM system. For anN-channel DWDM system, there are N transmitters 140 and N receivers 150with one transmitter 140 and one receiver 150 for each channel. Atransmitter 140 generates the optical carrier signal at the channelwavelength and modulates the carrier signal with a single data streambefore transmitting the modulated optical signal to the multiplexer 110.The multiplexer 110 then combines the N modulated optical signals havingdifferent channel wavelengths into a single multi-channel opticalsignal, and sends this through the fiber 120. The demultiplexer 130receives the multi-channel optical signal and separates it into thedifferent channel wavelengths. Each receiver 150 then demodulates one ofthe demultiplexed channel signals to extract the data signal. While FIG.1 shows a prior art simple system, it is understood that in real life, aduplex system is used, with one or more transmitters and receivers ateach end. Dutton, Harry J. R., Understanding Optical Communications,1998, pp. 513–568, ISBN 0-13-020141-3 presents a description of the DWDMsystem and of its components and is herein incorporated by reference.

The data rate (in bits per second or bps) through a single optical fibermay be increased by combining one or more of the following methods:increasing the data modulation rate; increasing the number of channelsper fiber; and selecting a modulation method having a higher spectralefficiency.

Increasing the data modulation rate is limited by semiconductortechnology and cost, as well as frequency-dependent fiber impairments aschromatic and Polarization Mode Dispersion (PMD). Increasing the numberof channels per fiber is limited by the properties of optical componentmaterials. Current and proposed implementations of DWDM systems use achannel modulation rate of about 10 GHz (OC-192) and use 40 channelsover the conventional optical band (C-band) between 1530 nm and 1560 nm.Therefore, the transmission bit-rate through a single optical fiber isabout 400 Gbps. Each channel has a bandwidth of about 100 GHz. Thespectral efficiency is defined as the channel bit-rate divided by thechannel bandwidth. The spectral efficiency of the system is therefore0.1 bit/Hz. The spectral efficiency may be doubled by using a coherentmodulation technique such as quadrature phase shift keying (QPSK). QPSKencodes two bits per modulation period and therefore doubles the channeltransmission bit-rate to 20 Gbps. The two bits encoded during QPSK arereferred to as a symbol and the modulation period is referred to as thesymbol period. The inverse of the symbol period is the symbol rate.

The channel bit-rate may also be doubled by combining two data streamsinto a single channel. U.S. Pat. No. 6,038,357 issued to Pan discloses afiber optic network that combines two data streams into a single channelby polarizing the optical signal modulated by the first data stream to apolarization plane that is orthogonal to the polarization plane of theoptical signal modulated by the second data stream.

Polarization mode dispersion (PMD) arises in optical fiber when circularsymmetry is broken by the presence of an elliptical core or bynoncircularly symmetric stresses. The loss of circular symmetry resultsin the difference in the group velocities associated with the twopolarization modes of the fiber. The main effect of the PMD is thesplitting of the narrow-band pulse into two orthogonally polarizedpulses (dual imaging) that propagate through the fiber with thedifferent group velocities. As the dual images propagate through thebirefringent fiber, there states of polarization (SOP) constantlyundergo changes causing the random coupling between the two images.

The PMD varies randomly from fiber to fiber. In the single fiber, thePMD also varies randomly with the optical carrier frequency and ambienttemperature. PMD broadens and degrades the signal and limits thedistance the signal may propagate before the information encoded in thesignal is lost.

Therefore, there remains a need to improve the spectral efficiency ofexisting/planned DWDM standards (OC-48 at 2.048 Gbps or OC-192 at 10Gbps) using existing fiber optic cables. There also remains a need forPMD compensation of the received optical signal.

SUMMARY OF THE INVENTION

In one aspect, the present invention is directed to an opticalcommunication system having a transmitter configured to receive a firstdata stream and transmit a code division multiplexed optical signalcomprising a number K code words modulated with first data from saiddata stream; and a receiver optically connected to the first transmitterand configured to receive said code division multiplexed optical signal,detect and demodulate said K code words within the code divisionmultiplexed optical signal, and output said first data, with the K codewords being orthogonal to one another. The code words in said codedivision multiplexed optical signal include H and V polarizations.

In another aspect, the present invention is directed to a node in anoptical communication system, the node having such a transmitterco-located with such a receiver.

In another aspect, the present invention is directed to a transmitterfor a code division multiplexed optical communication system for sendingK code words, each code having P pulses. In one embodiment, thetransmitter has a pulsed light source, a transmitter splitter configuredto output at least a number K identical code beams from the pulsed lightsource, a code modulator for each of the K identical code beams, eachcode modulator configured to receive one of the K identical code beamsand output a corresponding data beam, and a transmitter combinerreceiving and combining the K data beams to form a code divisionmultiplexed optical signal comprising K data-modulated code words.

In another embodiment, the transmitter of the present invention hasmultiple coding stages, including a first coding stage which employsdynamic code modulators and a second coding stage that employs pulsespreaders.

In another aspect, the present invention is directed to a code modulatorfor use in an optical transmitter. The code modulator includes a pulsespreader and a data modulator. The pulse spreader includes a splitterwhich splits an incoming pulse into a plurality of P parallel pulses, Pbeing the number of time chips in a code word, a time chip modulator todelay and code modulate each of the P pulses, and a P:1 combiner tocreate a single code word from the individually delayed and codemodulated pulses.

In another aspect, the present invention is directed to a pulse spreadercircuit configured to receive a single pulse within a predetermined timewindow and output an imprinted code beam having a number P modulatedpulses within that time window. A spreader calibration unit receives theimprinted code beam from the pulse spreader and a reference light sourceas inputs, and outputs a spreader control signal which is sent to thepulse spreader.

In another aspect, the present invention is directed to a spreadercalibration unit that ensures that each of the pulse spreaders hasproper modulation by detecting and correcting an error in phase among anumber P individual pulses belonging to one of a number K code words orpulse streams, each pulse occupying a chip period of length C.

In one embodiment, the optical pulse spreader calibration unit includesa first switch configured to select one from among (a) said K pulsestreams, and (b) a reference signal, to thereby output a first signal; asecond switch configured to select one from among (a) a pulsed lightsource having a pulse period of P*C and (b) to thereby output a secondsignal; a number P calibration delays each receiving the second signalas an input, the p-th calibration delay outputting a p-th delayedversion of the second signal delayed by (p−1)*C; a third switchreceiving and combining P delayed versions of the second signal from Pcorresponding calibration delays to thereby output a third signal; andan optical phase detector configured to receive the first and thirdsignals and output in-phase and quadrature signals derived from thefirst and third signals.

In another embodiment, the optical pulse spreader calibration unitincludes a first switch configured to select one of the K encoded pulsestreams and output a first signal; a phase detector having said firstsignal input thereto; a light beam directed into the phase detector viaa gate during a selected chip period determined by variable electronicdelay; wherein the phase detector determines an amplitude product of,and a phase difference between, the first signal and the gated lightbeam; and the amplitude product and phase difference are passed to aprocessor that determines an offset that should be applied to correct anerror in phase or amplitude of a pulse; and the processor outputs asignal to control the variable electronic delay.

In another aspect, the present invention is directed to an opticalsignal data modulator for use in a code modulator of the presentinvention. The optical signal data modulator is configured to receive anumber P time chips of a code word belonging to a family of orthogonalcodes, and modulate the orthogonal code words with data. The datamodulator includes a power splitter which splits the incoming pulsesinto H and V components, a pair of power splitters to power split eachcomponent into H1 and H2 and V1 and V2 sub-components, respectively, apair of phase shifters to phase shift the H2 and V2 sub-components by 90degrees from H1 and V1 accordingly, four bit-modulators to then modulateeach sub-component, a pair of first combiners to combine the twomodulated H components to form a data modulated H′ component and a datamodulated V′ component, and a polarization beam combiner which combinesthe data modulated H′ and V′ components into two orthogonalpolarizations.

In another aspect, the present invention is directed to a receiverhaving a receiver splitter configured to split a received informationsignal comprising K orthogonal data-modulated code words into Kidentical received information signals, and K code receivers, each codereceiver having a receiver pulse spreader configured to create areference signal which corresponds to one of the K orthogonaldata-modulated code words, and a receiver unit having the receivedinformation signal and the reference signal input thereto, wherein thereceiver unit is configured to detect and demodulate that one of the Kdata-modulated code words to which the reference signal corresponds.

In another aspect, the present invention is directed to an opticaldetection circuit for use in a receiver unit in accordance with thepresent invention. The optical detection circuit receives theinformation signal and the reference signal, and outputs in-phase andquadrature components of first and second orthogonal polarizationcomponents of the information signal. The optical detection circuitincludes a polarization beam splitter which splits the informationsignal into first and second orthogonal polarization components, a firstoptical phase detector that receives the first orthogonal polarizationcomponent and a reference signal as inputs, and outputs in-phase andquadrature components of the first orthogonal polarization component, asecond optical phase detector which receives the second orthogonalpolarization component and the same reference signal as inputs, andoutputs in-phase and quadrature components of the second orthogonalpolarization component, and a symbol synchronizer circuit that receivesthe in-phase and quadrature components of the first and secondorthogonal polarization components of the information signal, andoutputs at least one timing signal to synchronize symbol boundaries ofsaid data-modulated codewords in said optical phase detectors.

A polarization mode dispersion controller associated with the receiverreceives digitized in-phase and quadrature components of the first andsecond orthogonal polarization components of the information signal, andoutputs at least one polarization control signal in response thereto.

In another aspect, the present invention is directed to an optical phasedetector comprising an optical hybrid detector having first and secondsignal inputs and first and second signal outputs, the first signaloutput being proportional an in-phase difference between the first andsecond signal inputs, and the second signal output being proportional toa quadrature difference between the first and second signal inputs; afirst signal conditioning cascade circuit arranged to process the firstsignal output from the optical hybrid detector to thereby form adigitized in-phase component signal; and a second signal conditioningcascade circuit arranged to process the second signal output from theoptical hybrid detector to thereby form a digitized quadrature componentsignal.

In another aspect, the present invention is directed to an opticalhybrid detector having first and second signal inputs, and first andsecond signal outputs. The optical hybrid detector includes splitters,phase shifters and combiners to form the complex conjugate components ofthe first and second signal inputs. These components are input to a pairof matched detectors that output the in-phase and quadrature differencesbetween the first and second signal inputs.

In another aspect, the present invention is directed to a symbolsynchronizer for synchronizing symbols of a selected data signal from amultiplexed data signal. The symbol synchronizer uses thesymbol-to-symbol energy difference, instead of finding the maximumenergy.

In another aspect, the present invention is directed to a self-homodynereceiver. The self-homodyne receiver has a polarizing beam splitter tosplit an incoming optical code division multiplexed signal having Korthogonal code words modulated with data into first and secondorthogonal polarizations. The receiver has K sets of first and secondcode despreaders. The k-th set of the first and second despreaders isconfigured to output first and second phase information, respectively,of data modulated on the k-th code word having first and secondorthogonal polarizations, respectively. First and second optical phasedetectors compare the first/second phase information from a currentsymbol with the first/second phase information from an immediatelypreceding symbol, and output first/second signals reflective of thephase differences between data from the current and preceding symbols.

In another aspect, the present invention is directed to a PSP-based PMDcompensator, to compensate for PMD caused by the fiber and itsenvironment over a long distance run, without the use of a repeater. Thereceiver restores a transmitted signal's polarization by aligning theSOP of the received signal with the SOP of the signal launched into thefiber. The transmitter selects the optimal SOP to launch the opticalsignal into the fiber by aligning the SOP with the principal state ofpolarization (PSP) axes of the fiber. The SOP transmitted along the PSPaxes exhibits the least frequency dependence that minimizes the couplingof the propagating signal. The polarization transformation in the fiberis a time-varying process; therefore, both the receiver PMD compensatorand the transmitter SOP compensator must be able to track those changes.The receiver sends back the information to the transmitter via aseparate channel to adjust the transmitter SOP compensator to re-aligntransmitted SOP with PSP.

In yet another aspect, the present invention is directed to a method foradjusting the receiver-end polarization compensation device, and methodfor adjusting a transmitter-end polarization compensation device.

BRIEF DESCRIPTION OF THE FIGURES

The present invention may be understood by reference to the followingdetailed description of the preferred embodiment of the presentinvention, illustrative examples of specific embodiments of theinvention and the appended figures in which:

FIG. 1 is a block diagram of a simplex prior art DWDM systemarchitecture;

FIG. 2 a is a block diagram of a simplex optical communication system inaccordance with a preferred embodiment of the present invention;

FIG. 2 b is a block diagram of one-half of a duplex opticalcommunication system in accordance with a preferred embodiment of thepresent invention;

FIG. 3 is a block diagram of a transmitter in accordance with oneembodiment of the present invention;

FIG. 4 is a block diagram of the pulse spreader used in the transmitterof FIG. 3;

FIG. 5 is a block diagram of a first embodiment of a spreader calibratorused with the spreader of FIG. 4;

FIG. 6 is a block diagram of a second embodiment of a spreadercalibrator;

FIG. 7 is a block diagram of the transmitter-side optical phase detectorin a preferred embodiment of the present invention;

FIG. 8 is a block diagram of the data modulator in a preferredembodiment of the present invention;

FIG. 9 a is a block diagram of the channel receiver in accordance withone embodiment of the present invention;

FIG. 9 b is a block diagram of the channel receiver in accordance withanother embodiment of the present invention;

FIG. 10 is a block diagram of the data demodulator in a preferredembodiment of the present invention;

FIG. 10 a depicts one configuration for a PMD compensator/digital datademodulator circuit;

FIG. 10 b depicts a second configuration for a PMD compensator/digitaldata demodulator circuit;

FIG. 11 is a block diagram of the recevier-side optical phase detectorin a preferred embodiment of the present invention;

FIG. 12 is a block diagram of the symbol synchronizer in a preferredembodiment of the present invention;

FIG. 13 a is a graph of the average symbol energy as a function of thesymbol time;

FIG. 13 b is a plot of the average symbol-to-symbol energy difference asa function of the symbol time;

FIG. 14 a is a block diagram of the transmitter-receiver link of thepresent invention;

FIG. 14 b is a flow diagram of a PMD controller for compensating thereceiver of FIG. 14 a;

FIG. 14 c is a flow diagram of a PMD controller for compensating thereceiver and the transmitter of FIG. 14 a; and

FIG. 15 is a block diagram of a transmitter having multiple codingstages.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 2 a is a block diagram of the system architecture of a simplexcommunication system 249 in accordance with the present invention. Thecommunication system 249 has a transmitter end 259 and a receiver end269. The transmitter end 259 has N transmitter modules 257, eachcorresponding to one of N channels, each channel being characterized bya different wavelength, λ_(i), where the index, i, runs from 1 to N. Thereceiver end has N corresponding receiver modules 267. As seen in FIG. 2a, each transmitter module 257 includes a transmitter 250 and preferablyalso incorporates transmitter-end state of polarization (SOP)compensation 275 positioned between the transmitter and the multiplexer.One such transmitter-end SOP compensator is provided for each channel.Similarly, each receiver module 267 includes a receiver 260 andpreferably also incorporates a receiver-end polarization mode distortion(PMD) compensation 285. As seen in FIG. 2 a, the receiver PMDcompensation is positioned between the demultiplexer and each receiver,with one such compensator again being provided for each channel.

Each transmitter module 257 receives, for its channel, a total of K datastreams, the data streams being depicted by DATA_(ijk), where subscript‘i’ is again the channel index, j is the polarization index (either ‘H’or ‘V’) and subscript ‘k’ is the data stream index. For a given ‘i’ and‘k’, the ‘H’ and ‘V’ data streams are preferably used to modulate thesame code word on H and V polarizations. Thus, each code word, whentransmitted, is used on both polarizations, and each polarizationcarries different data DATA_(iHk) and DATA_(iVk). Thus, since the index‘j’ takes on only two possible values, one may consider the incomingdata streams to effectively comprise 2K data streams, each having afirst component that will ultimately be encoded and sent on Hpolarization and a second component that will ultimately be encoded andsent on V polarization.

While the above ‘data stream’ nomenclature may be expedient toillustrate the effects of coding in accordance with the presentinvention, it should be kept in mind that a single data stream may beformatted in any number of ways including the “DATA_(ijk)”representation discussed above. What is important is that the incomingdata preferably is used to modulate the same code word on both H and Vpolarizations, with a total of K code words being combined andtransmitted on each channel. Each code word includes both the ‘H’ and‘V’ data components, i.e., both ‘j’ values from a “DATA_(ijk)” datastream and these are encoded in a single code word.

Each transmitter module 257 preferably encodes the K data streams usingK codes created by its transmitter 250, and preferably four bits areencoded on each code word. The encoded signal for each channel is sentto a multiplexer 210 which combines the encoded channels into a singleencoded multi-channel optical signal. The encoded multi-channel opticalsignal is sent over an optical fiber 220 to a demultiplexer 230 whichseparates the encoded multi-channel optical signal into the variouschannel wavelengths. The individual channel wavelengths are then sent ona receiver module 267 corresponding to that channel. If desired, opticalamplifiers 222 may be used, although this is not a requirement of thepresent invention. At the receiver module 267, the received signal isreceived and demodulated by the receiver 260 which outputs the K datastreams.

When compensation is provided for, the receiver circuitry 260 sends anauxiliary signal 295 to the transmitter-side SOP compensator 275. Thispreferably is done out-of-band, on a continuous basis to mitigatepolarization mode distortion caused by the fiber and the other optics.

FIG. 2 b shows a node 299 representing one-half of a bidirectional fiberoptic communication link in accordance with the present invention. Thenode 299 preferably accommodates a number N of channels 240, eachchannel comprising a set including a transmitter 250 and a receiver 260which are co-located. Indeed, the entire suite of all N channelspreferably are placed at a single site or node.

Each end of the link has a DWDM multiplexer (Mux) 210 connected to thedownstream optical fiber 220 and a DWDM demultiplexer (Demux) 230connected to the upstream optical fiber 225. The downstream fiber 220 isconnected to the Demux at the other end of the link and the upstreamfiber 225 is connected to the Mux at the other end of the link. In apreferred embodiment, the optical fibers 220 225 are single mode fiberscapable of transmitting optical signals in the C-band (wavelengthsbetween 1530 nm and 1560 nm). The Mux 210 multiplexes N channels 240 fortransmission through the fiber 220 and the Demux 230 demultiplexes thereceived optical signal from the fiber 220 into N channels 240. Forpurposes of clarity, FIG. 2 shows only two channels 240 but it isunderstood that N channels are represented in FIG. 2.

Each channel 240 is characterized by a different wavelength, λ_(i),where the index, i, runs from 1 to N. In a preferred embodiment, N isequal to 40 thereby giving each channel a bandwidth of about 100 GHz.Each channel 240 is capable of carrying K codes 280 and each code 280carries a data stream. For each channel 240, a transmitter (Tx) 250creates the K codes, modulates each code 280 with a data stream, andcombines the K data modulated codes into an optical channel signal 245before passing the optical signal 245 to the multiplexer 210. Eachchannel 240 also includes a receiver (Rx) 260 that recovers the K datastreams from the Demux 230.

In a preferred embodiment, K=16 codes are carried in each of the N=40channels of the C-band at a symbol rate of 2.5 GHz with a symbol periodof 400 ps, and 4 bits per symbol. This results in a capacity of 6.4 Tb/sand a spectral efficiency of 1.6 bits/Hz. Each code is further timesliced into 16 chips per symbol period for a chip period C=25 ps.

The detailed description of each component is now described. The Mux210, Demux 230, optical fiber 220, and fiber amplifier components 222are known to one of skill in the optical communications art and need notbe described further. Common devices such as splitters, combiners, phaseshifters, optical switches, delay lines, intensity and phase modulatorsand CW lasers are also known to one of skill in the opticalcommunication art and need not be described further. Similarly, oneskilled in the art of digital signal processing is able to implement thefunctions of adders, subtractors, squaring and absolute value using aDSP chip and also designing amplifiers and other standard componentswithout undue experimentation. Accordingly details of suchimplementations are not included herein.

Transmitter

FIG. 3 is a block diagram of a transmitter 250 configured to create acode division multiplexed signal in accordance one embodiment of thepresent invention. Coherent pulsed light source 310 generates a linearlypolarized monochromatic light beam characterized by wavelength, λ_(i).In a preferred embodiment, the coherent pulsed light source 310comprises a continuous wave laser 311 that generates a coherentmonochromatic light beam. The light beam is directed into an intensitymodulator 312. The intensity modulator 312 is, in turn, controlled by apulse generator 313. In a preferred embodiment, the pulse generator 313and intensity modulator 312 produce an optical stream having a pulseperiod (time between pulses) of T=400 ps, which corresponds to a 2.5 GHzsymbol frequency, and a pulse width between 10 ps and 20 ps. In anotherembodiment, the coherent light source 310 is a mode-locked laser thatdirectly generates a similar pulse stream.

The pulsed light beam 318 is passed through a splitter 320 that splitsthe light beam 318 into K code beams 322. Each code beam 322 is modifiedto produce K unique codes using a code modulator 360. For purposes ofclarity, FIG. 3 shows the modification of only one of the K code beams322 and it is understood that the remaining code beams are directed tostructures identical to those designated as 360 in FIG. 3. In apreferred embodiment, K is equal to 16 although any number of code beams322 may be used and is within the scope of the present invention.

Each code beam 322 is passed through pulse spreader 330 wherein eachcode beam 322 is imprinted with a unique code comprising P pulses andhaving duration T. Each imprinted code beam 332 is then passed through adata modulator 340. The data modulator 340 modulates the imprinted codebeam 332 with a data stream 370 to produce a data beam 342. Data beam342 comprises two orthogonal polarizations, each modulated independentlyby the data, as discussed below with regard to the data modulator 340.Each data beam 342 is then combined with the other data beams havingwavelength λ_(i) in a combiner 350 to form a channel beam. The channelbeam output by the transmiter is thus a code division multiplexedoptical signal comprising K data-modulated code words which are sent onthe channel 240. A single channel 240 is therefore capable of carrying Kdata streams.

In a preferred embodiment, a single spreader calibration unit 335controls each of the K pulse spreaders 330 in the i-th channel only. Forthis, the spreader calibrator 335 monitors each of the K imprinted codebeams 332 and controls the K pulse spreaders 330 with a spreader controlsignal 336 to insure the accuracy and stability of each of the imprintedcode beams 332. In another embodiment, a spreader calibration unit 335controls each of the K pulse spreaders 330 for all N channels.

Pulse Spreader

FIG. 4 is a block diagram of a preferred embodiment of the pulsespreader 330. A portion of the pulsed light beam 318 is directed to thespreader calibration unit 335 while the code beam 322 is split by a 1:Psplitter 410, P representing the number of code bits (referred to a“chip”) in each code word. In a preferred embodiment, the code beam issplit into P=16 split beams, however, any number of split beams may beused and is within the scope of the invention. For purposes of clarity,FIG. 4 shows the modification of only one of the P split beams and it isunderstood that the remaining split beams are directed to structuresidentical to the chip modulators 450 of FIG. 4.

Each of the P split beams is passed through a delay 420 that delays thesplit beam by a predetermined amount. In a preferred embodiment, thedelay 420 is a multiple of TIP where T is the symbol period and P is thenumber of split beams or chips. In a preferred embodiment, the delay 420may be a calibrated length of waveguide such as an optical fiber or asemiconductor waveguide. The delay increment, C=TIP, is referred to as achip period and there are P chip periods per symbol period. In apreferred embodiment, a symbol period of T=400 ps is selected and P=16is chosen, resulting in a chip period of C=25 ps.

The first split beam p=1 is not delayed, the second split beam p=2 isdelayed by one chip period, the third split beam is delayed by two chipperiods p=3, etc., and the pth split beam is delayed by p=(P−1) chipperiods. The result of the family of delays 420 acting on the P splitbeams provides each split beam with a single pulse (preferably of length10–20 ps), each pulse being positioned in a unique “time chip” withinthe symbol period T, the time chips being spaced apart in time by T/P.If the split beams are now combined, the repetition rate of the pulsetrain of the combined beam would be multiplied by a factor of P, or inthe case of the instant example, would increase from 2.5 GHz to 40 GHz.

The split beams, however, are not combined immediately after the delays420. Instead, each delayed split beam is passed through a chipmodulation circuit 430. In a preferred embodiment, the chip modulationcircuit 430 phase shifts each pulse by a predetermined amount, δ_(kp),where δ_(kp) is the phase shift applied to the p-th chip of the k-thcode. In a preferred embodiment, the chip modulation circuit 430 is anopto-electronic device such as a Pockels cell phase modulator. Inanother embodiment, the chip modulation circuit 430 may also apply anamplitude scaling, a_(kp), to each split beam. Regardless of how thechip modulation circuit 430 is implemented, combiner 440 combines eachof the delayed-and-now-chip-modulated split beams into a single codebeam 332. A portion of the code beam 332 exiting the combiner 440 isdirected to the spreader calibration unit 335.

The set of modulations provided by the chip modulator, {δ_(k1), . . . ,δ_(kP); a_(k1), a_(kP)}, uniquely identify the k-th code and are chosento be orthogonal to each of the other (K−1) codes. The selection of theorthogonal code sets may be from any one of the orthogonal code setsknown to one of skill in the signal processing arts. In a preferredembodiment, the modulation consists of phase shifts given by thefollowing equation:

$\begin{matrix}{\delta_{kp} = \frac{2{\pi\left( {k - 1} \right)}\left( {p - 1} \right)}{P}} & (1)\end{matrix}$

-   -   where p is the chip index varying from 1 to P where P is the        number of chips per symbol period and k is the code index        varying from 1 to K where K is the number of codes per channel.        In the preferred embodiment, K≦P.

Each chip modulation circuit 430 receives input from the spreadercalibration unit 335 via a driver 435 belonging to the chip modulator450. The spreader calibration unit 335 determines the amount of phaseshift and/or amplitude scaling each chip modulation circuit 430 appliesto its respective split beam. The spreader calibration unit 335 adjustseach shift/scaling by a calibration offset that it determines. Thespreader calibration unit 335 may be implemented in a digital signalprocessor (DSP) of the sort known to those skilled in the digitalprocessing art. In a preferred embodiment, a single spreader calibrationunit 335 controls the P chip modulating circuits 430 for all K codes. Aswill be clear to one of skill in the art, one may use multiple spreadercalibration units, such as one for each of the K codes.

The chip modulation circuit 430 is essentially a static modulatorbecause the modulation applied to each chip does not vary except forlong term (with respect to the symbol period) drifts or slowly varyingconditions such as temperature that are compensated by the spreadercalibrator 335. By splitting the code beam 322 into P chips andindependently modulating each chip individually every symbol period, thek-th pulse spreader 330 creates an imprinted code beam that carries thecode beam's identification at a modulation rate of P times the symbolrate using inexpensive (relative to dynamic modulators capable ofmodulating at P times the symbol rate) static modulators. However, analternative to the pulse spreader 330 described above with respect toFIG. 4, is to employ a pulse spreader having a dynamic code modulatorcapable of modulating the code beam 322 at P times the symbol rate, andthis is also contemplated in the system of the present invention.

FIG. 5 is a block diagram of a preferred embodiment of the spreadercalibration unit 335. The spreader calibration unit 335 adjusts themodulation applied by the chip modulation circuit 430 to ensure the Kcode beams remain orthogonal to each other. A portion of the imprintedcode beam (ICB) 332 from each of the K pulse spreaders 330 is directedinto an ICB switch 510. In a preferred embodiment, a portion of thepulsed light beam 318 is directed into a CB switch 520. Acontinuous-wave (CW) reference laser 530 generates a reference beam thatis directed to both the ICB switch 510 and the CB switch 520 by splitter532. In an alternate embodiment, a portion of the K code beams (CB) 322may be directed into the CB switch 520.

At any instant, the ICB switch 510 selects one from among the K+1 inputsand directs the ICB-selected beam 512 to an optical phase detector 550.In other words, the ICB switch 510 selects either one of the K beams 332from pulse spreader 330 or the reference beam and sends the selectedbeam onwards.

At any instant, the CB selects either the reference laser beam, or thepulse light source beam. The CB switch 520 directs the selected beam to1:P splitter 524 which splits the selected beam into P split beams. Eachsplit beam is passed through a unique delay line 525. Each delay line525 adds a known delay to the split beam that corresponds to the delayof one of the P chips. For example, the p-th delay line will correspondto the p-th chip in each symbol period. The P split beams are directedinto a calibrated chip switch 540. The calibrated chip switch 540selects one of the P delayed split beams and directs the selecteddelayed split beam 522 into the optical phase detector 550. The opticalphase detector 550 calculates the (complex) product of the ICB-selectedbeam 512 and the complex conjugate of the selected split beam 522,giving the product of their amplitude and the phase difference betweenthe two beams.

When the imprinted code beam 332 for code k is selected by ICB switch510, the p-th chip of the ICB-selected beam 512 will be proportional toa_(kp)·e^(j2πf(t−τ) ^(kp) ⁾·e^(jδ) ^(kp) , where f is the known beamfrequency and δ_(kp), a_(kp) and τ_(kp) are the preset phase shift,amplitude scaling and delay applied for the p-th chip by the k-th pulsespreader 330. When the pulsed light beam 318 is selected by CB switch520, the selected split beam 522 will have a zero signal for all chipsexcept for the chip corresponding to the selected delay line 525. Forexample, if the split beam switch 540 selects the second delay linecorresponding to the second chip of the symbol period, the selectedsplit beam will have a zero signal for the first chip period, a non-zerowaveform in the second chip period, and zero signals for the thirdthrough P-th chip periods. Selected split beam 522 will be proportionalto e^(j2πf(t−τ) ^(p) ⁾ where τ_(p) is the known p-th delay selected byswitch 540.

Since the selected split beam 522 has a non-zero signal only for thep-th delay line (selected by the split beam switch 540), the productdetermined by the optical phase detector 550 is equal toa_(kp)·e^(j2πf(τ) ^(p) ^(−τ) ^(kp) ⁾·e^(jδ) ^(kp) . Processor 560measures the amplitude a_(kp) and the phase δ_(kp)−2πf(τ_(kp)−τ_(p)) andcommands the spreader calibration unit 335, for the k-th pulse spreader,to maintain a_(kp) and δ_(kp)−2πfτ_(kp) at the required value given, forexample, by equation 1.

Each of the P delay lines 525 (τ_(p)) are calibrated using the referenceCW laser 530 (which may be tapped from CW laser 311) as will now bedescribed. The ICB switch 510 selects the reference laser beam anddirects the beam to the optical phase detector 550. CB switch 520selects the reference laser beam and directs the reference beam to the1:P splifter 524 where the reference beam is split into P split beams.Each split beam is directed to a delay line 525. A calibrated chipswitch 540 selects one of the delayed split beams and directs theselected delayed split beam 522 to the optical phase detector 550. Theoptical phase detector 550 determines the phase difference between thereference beam selected by the ICB switch 510 and the delayed referencebeam selected by the split beam switch 540. The reference beam isproportional to e^(j2πf) ⁰ ^(t) where f_(o) is the reference laser'sknown frequency and the delayed reference beam is proportional toe^(jπf(t−τ) ^(p) ⁾. The phase difference between the two referencebeams, 2πf₀τ_(p), gives the delay of the selected delay line 525.Processor 560 stores the phase differences for each of the delay lines525. Processor 560 also controls the switching of the ICB switch 510,the CB switch 520, and the split beam switch 540.

It is understood in the above description of the spreader calibrationcircuit 335 of FIG. 5 that the various switches 510, 520 and 540 arecontrolled by circuitry that has been omitted in the figure. In general,either processor 560, or another controller, or the like, will governthe timing and selection of these switches.

FIG. 6 is a block diagram of another embodiment of a spreadercalibration unit. In spreader calibration unit 335 a, an imprinted codebeam (ICB) switch 610 selects one of the K imprinted code beams from thepulse spreader 330 and directs the selected imprinted code beam 612 intoan optical phase detector 650. A continuous-wave (CW) reference laser620 generates a reference light beam 622 that is directed into the phasedetector 650 after gate 630. The gate may be implemented by aMach-Zehnder intensity modulator or the like. The gate 630 allows thereference light beam 622 to pass through the gate 630 to the phasedetector 650 only during a selected chip period. The selected chipperiod is determined by variable electronic delay 635, which iscontrolled by processor 660. The phase detector 650 determines theamplitude product of, and the phase difference between, the selectedimprinted code beam 612 and the gated reference beam 632. The amplitudeproduct and phase difference are passed to a processor 660 thatdetermines the offset that should be applied by the (kp)-th chipmodulation circuit.

FIG. 7 presents a block diagram of the optical phase detector 550, 650,seen in FIGS. 5 and 6. The optical phase detector receives two opticalbeams 705, 701 and generates two electrical signals 795, 791corresponding to the in-phase and quadrature components of themultiplication of 705 by the complex conjugate of 701. One of the twobeams, designated as A in FIG. 7, is referred to as a signal beam. Thesecond optical beam 701, designated as B in FIG. 7, is referred to forconvenience as a reference beam in this description of the optical phasedetector. It should be kept in mind, however, that the ‘reference beam’701 may be something other than an unmodulated train of pulses having noinformation.

The signal beam 705 is split into four beams identified by 715 in FIG.7. In a referred embodiment, the signal beam 705 is split into fourbeams by a cascade of 1:2 splitters 710. In another embodiment, thesignal beam 705 may be directly split into four beams by a single 1:4splitter. Regardless of how they are formed, the four signal beams 715are directed into four 2:1 combiners 730 a, 730 b, 730 c, and 730 d.

The reference beam 701 is first subjected to a 1:2 splitter 720 a toform identical beams R1 and R2. Beam R1 is then subjected to a second1:2 splitter 720 b to form identical beams R3 and R4. Meanwhile, Beam R2is first subjected to a 180° phase shifter 724 before being spilt by 1:2splitter 720 c to thereby form identical beams R5 and R6. Beam R3 isinput as signal 721 to combiner 730 a while beam R4 is first subjectedto a first 90° phase shifter 722 a before being input to combiner 730 bas signal 723. Beam R5 is input as signal 725 to combiner 730 c whilebeam R4 is first subjected to a second 90° phase shifter 722 b beforebeing input to combiner 730 d as signal 727.

The resulting beams R3, R4, R5 and R6, represented as signals 721, 723,725, and 727 are phase-shifted by 90° increments. Beam R3/721 has zerophase shift and is combined with one of the four signal beams 715 in 2:1combiner 730 a to produce first combined beam 731 that is the sum of thesignal and reference beam, designated as A+B. Beam R4/723 has a 90°phase shift and is combined with one of the four signal beams 715 insecond 2:1 combiner 730 b to produce second combined beam 733 that isdesignated as A+jB. Beam R5/725 has a 180° phase shift and is combinedwith one of the four signal beams 715 in third 2:1 combiner 730 c toproduce third combined beam 735 that is the difference between thesignal beam and reference beam and is designated as A−B. Finally, beamR6/727 has a 270° phase shift and is combined with one of the foursignal beams 715 in fourth combiner 730 d to produce fourth combinedbeam 737 that is designated as A−jB.

The first and third combined beams 731, 735 are input to a first matcheddetector 740 a. The first matched detector 740 a includes light sensors745 a, to thereby generate electrical signals 798 a that areproportional to the intensity difference between the first and thirdcombined beams 731, 735. The light sensors 745 a are preferablyphotoelectric detectors such as p-n, p-i-n, or Schottky-barrierphotodiodes, and are selected to generate substantially identicalelectrical signals for the same incident light beam. The electricalsignals 798 a generated by the first matched detector 740 a are input toa first amplifier 750 a. The output signal 752 of the first amplifier750 a is proportional to the in-phase difference between the signal beam705 and the reference beam 170 (real{AB*}).

The second and fourth combined beams 733, 737 are directed to a secondmatched detector 740 b comprising matched light sensors 745 b, tothereby generate electrical signals that are proportional to theintensity difference between the second and fourth beams 733, 737. Theelectrical signals 798 b generated by the second matched detector 740 bare input into a second amplifier 750 b. The output signal 754 of thesecond amplifier 750 is proportional to the quadrature phase differencebetween the signal beam 705 and the reference beam 701 (imag{AB*}). Thesplitters, phase shifters, combiners and detectors together comprise anoptical 90° hybrid detector 799 that outputs the signals 798 a, 798 b.

The analog output from the amplifiers 750 a, 750 b are input to low passfilters 760 a, 760 b, respectively. The output of the lowpass filters isthen subject to dc bias adjustment 770 a, 770 b based on bias signals772, 773, respectively, provided by a controller (not shown) whichestimates signal energy. The filtered and bias-adjusted analog signalsI′ and Q′ on lines 775, 777, respectively, are converted to digital formby sample and hold (“S&H”) units 780 a, 780 b, respectively, which aretimed by synchronization input 781. The output of the sample and holdunits is sent on to analog-to-digital converters 790 a, 790 b, to formthe in-phase I 795 and quadrature Q 791 signals which are subject tofurther processing, as is known to those skilled in the art.

Data Modulator

FIG. 8 is a block diagram of a preferred embodiment of a data modulator340 in accordance with the present invention. The imprinted code beam332 from pulse spreader 330 enters the data modulator 340 through apower splitter 810. The power splitter 810 splits the imprinted codebeam 332 into two beams, designated as H and V. The H and V beams areeach directed into separate 1:2 splitters 820 a, 820 b, although a 1:4splitter may be used instead. The splitter 820 a splits the H beam intoH1 and H2 components and directs one of these (H2 in FIG. 8), into a 90°phase shifter 825 a. Similarly, the splitter 820 b splits the V beaminto V1 and V2 components and directs one (V2 in FIG. 8) into a 90°phase shifter 825 b.

Each of the four component beams are directed into a separate modulator830 a, 830 b, 830 c, 830 d where the data stream 370 is modulated on thefour component beams. As discussed above, the data stream DATA_(ijk)includes data to be encoded on the H and V polarizations of a singlecode word, depending on the index ‘j’. Thus, the data input tomodulators 830 a, 830 b is the data corresponding to j=‘H’ and the datainput to modulators 830 c, 830 d is the data corresponding to j=‘V’.

Control signals from a power balancer 345 ensure that the intensity ofeach data-modulated component beam is comparable to that of the othersand ensures the orthogonality of the in-phase and quadrature componentsof the constellation. Thus, the power balancer may have inputs into oneor both of the phase shifters 825 a, 825 b and the data modulators 830a, 830 b, 830 c, 830 d.

In a preferred embodiment, each of the four component beams is modulatedwith one bit. The modulators 830 a, 830 b, 830 c, 830 d are eachimplemented as a Mach-Zehnder Interferometer (MZI), each applying anamplitude of 1 or −1 (00 or 180° phase shift) to the component beamdepending on whether the bit state is a 0 or 1. In a preferredembodiment, the data is encoded in the changes of phase between one bitto the other, referred to as differential quadrature phase shift keying(DQPSK). As will be clear to one of skill in the art, the 90° shift maybe applied after the bit modulator 830. As will also be clear to one ofskill in the signal processing arts, the modulator 830 may also be aphase modulator or a multilevel amplitude modulator thereby allowing anyamplitude/phase modulation and encoding of a higher number of bits persymbol.

The modulated H1 and H2 component beams are combined in a first combiner840 a to form a data-modulated beam H′. Similarly, the modulated V1 andV2 component beams are combined in a second combiner 840 b to form adata-modulated beam V′. The data-modulated beams H′ and V′ are thencombined in a polarization beam combiner 850 that rotates thepolarization plane of the H′-beam 90° to the polarization plane of theV′-beam and combines both beams into the data beam 342. The combined H′and V′ beams do not interfere because of their orthogonal polarizationplanes.

As will be clear to one of skill in the optical arts, the polarizationstate of H′ and V′ beams may also be converted to any two orthogonalpolarizations (such as right circularly polarized beam and a leftcircularly polarized beam) before combining into the data beam and stillmaintain the orthogonality condition that prevents interference betweenthe two beams.

Receiver

FIG. 9 a is a block diagram of one embodiment of channel receiver 260 inaccordance with the present invention. After demultiplexing by the DWDMdemultiplexer 230, each channel beam, indicated by λ_(i) in FIG. 9 a, isdirected into its own channel receiver 260 where the K data streams arerecovered and converted into electrical signals for downstreamprocessing. The channel beam first is directed into an Rx PMDcompensator 910. The Rx PMD compensator 910 adjusts the polarizationstate of the channel beam to compensate for PMD distortions to thechannel beam during transmission of the channel signal through theoptical fiber. In a preferred embodiment, the Rx PMD compensator 910 isthe Acrobat™ Polarization Control Module (PCM) from Corning Incorporatedof Corning, N.Y. The PolarRITE™II Polarization Controller from GeneralPhotonics Corporation of Chino, Calif. may also be used as the Rx PMDcompensator 910. As seen in FIG. 9 a, the Rx PMD compensator 910receives controls signals 1055 from the receiver unit circuitry 930which performs the functions of data extraction and PMD control, asdiscussed further below.

The compensated channel beam is directed into a 1:K splitter 920. Thesplitter 920 splits the compensated channel beam into K split beams 925.Each of the K split beams 925 are directed into a code receiver 960.Each code receiver 960 is configured to recover one of the Kdata-modulated codes from the split beam 925 and convert thedata-modulated code into an electrical signal corresponding to the datastream 932. For purposes of clarity, only one code receiver is shown inFIG. 9 a and it should be understood that the channel receiver 260 has aseparate code receiver 960 for each of the K data streams in the channelsignal.

Pulsed light source 940 is substantially similar to pulsed light source310 at the transmitter, and can actually be tapped from source 310 inthe “duplex” node arrangement of FIG. 2 b. Pulse light source 940generates a reference optical signal, characterized by λ_(i). Thereference optical signal is directed into the receiver pulse spreader950 of each of the code receivers 960.

The structure and operation of the receiver pulse spreader 950 issubstantially the same as the pulse spreader 330 described with respectto FIG. 4 with the amplitude scaling 1/a_(kp) instead of a_(kp). Thus,the receiver pulse spreader 950 has an associated calibration unit (notshown) which operates in a manner similar to that described with respectto spreader calibration units 335 or 335 a. The receiver pulse spreader950 modulates the reference optical signal to produce a referenceimprinted code beam 955 corresponding to one of the K imprinted codebeams used to carry the K data steams in the channel signal. Thereference imprinted code beam 955 is directed into the receiver unit930, and the optical detection circuitry 1090 and the demodulationcircuitry 1060 within the receiver unit 930 recovers the data stream 932from the channel signal 925 corresponding to the reference imprintedcode beam 955.

The PMD controller portion 1050, 1050 b within receiver unit 930, has anassociated DSP which outputs a PMD compensator signal 1055 which is fedback to the Rx PMD compensator 910 (depicted in FIGS. 2 a & 2 b as‘280’), and a SOP compensator signal 1056 which is sent to thetransmitter over auxiliary channel 290.

Receiver Unit

FIG. 10 is a block diagram of one embodiment of the receiver unit 930seen in FIG. 9 a. This unit 930 employs homodyne detection. A split beamsignal 925 is directed into a polarizing beam splitter 1010 of theoptical detection circuitry 1090. The polarizing beam splitter 1010splits the split beam 925 into its first and second orthogonalpolarization components, designated as H and V in FIG. 10.

The H-beam 1012 is directed to a variable delay 1015 controlled by adelay signal 1014 from a symbol synchronizer 1040. The variable delay1015 delays the H-beam to synchronize the H-beam to the V-beam 1011 sothat the H-beam and V-beam comprising the same symbol appear at thedemodulator at the same time (alternatively, the inverse delay can beapplied to the V-beam). The delayed H-beam is directed into a firstoptical phase detector 1030 a. The V-beam 1011 is directed into a secondoptical phase detector 1030 b. Reference imprinted code beam 955corresponding to one of the K codes entering the receiver unit 930 issplit by a 1:2 power splitter 1020. The split reference code beams 1022a, 1022 b are directed into optical phase detectors 1030 a, 1030 b,respectively.

Optical phase detector 1030 a generates a pair of digital signals H_(i),H_(Q), which represent the in-phase and quadrature components of the Hbeam. Optical phase detector 1030 a also generates a pair of analogsignals H′_(i), H′_(Q), which represent the analog versions of H_(i),H_(Q) from which the digital signals were formed. Similarly, opticalphase detector 1030 b generates a corresponding pair of digital signalsV_(i), V_(Q), which represent the in-phase and quadrature components ofthe V beam, and the associated analog signals V′_(i), V′_(Q). The timingof the phase detectors 1030 a, 1030 b are controlled by a timing signal1242 from the symbol synchronizer 1040.

The four analog signals H′_(i), H′_(Q), V′_(i), V′_(Q) are input intothe symbol synchronizer 1040 and the digital signals H_(i), H_(Q),V_(i), V_(Q), are input into the PMD controller/digital data demodulatorcircuit 1045. The data and the PMD control signals are ultimatelyextracted from the digital signals, while symbol synchronization isperformed using the analog signals.

The PMD controller 1050 receives the digitized I and Q components of theH and V polarizations and output the polarization control signals inresponse thereto. The PMD controller 1050 (see FIG. 10 a) determines theamount of PMD the optical signal has experienced and provides acompensation signal 1055 to the PMD compensator 910. The four signalsare further digitally processed in the digital data demodulator 1060 torecover the data stream 932. As will be clear to one of skill in thesignal processing art, the digital data demodulator 1060 may includecomponents such as an equalizer, differential demodulation, frequencycompensation and symbol timing recovery. In a preferred embodiment, thecomponents are implemented as algorithms/computer programs executing ona DSP, FPGA, ASIC or other processor known, to one of skill in thesignal processing art.

FIG. 10 a shows a conceptual configuration of data flow in a PMDcontroller/digital data demodulator circuit 1045 used in conjunctionwith a receiver having a receive signal PMD compensator 910 at thereceiver front end, such as seen in the receiver 260 of FIG. 9 a. Asdiscussed above, the optical detection circuitry 1090 outputs the I andQ components of the H and V polarizations of the incoming signal 925. InFIG. 10 a, the output S of the optical detection circuitry 1090represents the digital signals H_(i), H_(Q), V_(i), V_(Q), which havealready been PMD compensated. These four signals are input into both thedigital data demodulator 1060 and also to the digital PMD controller1050. The PMD controller, in turn, outputs a PMD controller signal 1055which is fed to the receiver's PMD compensator 910, and also outputs aSOP controller signal 1056 which is fed back to the transmitter's SOPcompensator. Thus, the shown digital polarization compensator systemprovides feedback to both the receiver input and the transmitter output.And, as stated above, the digital PMD controller 1050 and the digitaldata demodulator 1060 are preferably implemented as algorithms incomputer software.

FIG. 10 b shows a conceptual configuration of data flow in a PMDcontroller/digital data demodulator circuit 1045 of the sort used with areceiver having back-end digital PMD compensation 1050 c, instead of aninput PMD compensator 910, as seen in FIG. 9 a. In this instance, theuncompensated digital output S′ of the optical detection circuitry 1090is first subject to digital PMD compensation to thereby produce a PMDcompensated digital signal S. The signal S is then input to a digitalPMD compensation controller 1050 b which performs the necessarycalculations and adjustments to provide the needed digital PMDcompensation. It is understood that the configuration in FIG. 10 b isused with a receiver not having a front-end PMD compensator, and so nosignal 1055 is provided to a compensator device such as theaforementioned Acrobat™ Polarization Control Module (PCM). However, inthe configuration of FIG. 10 b, one may optionally still have a signal1056 b that is sent back to the transmitter 250 for transmitter-end SOPcompensation.

FIG. 9 b is a block diagram of another embodiment of a channel receiversuitable for use with the present invention. Channel receiver 260 aemploys self-homodyne detection, thereby obviating a need for a pulsedlight source 940 of the sort used in channel receiver 206.

The incoming channel beam 925 is first input to a receiver-end PMDcompensator 1105 and resulting compensated signal is input to a 1:Ksplitter 1107 to create K identical compensated signals, one for eachcode word. Each of these is then split by a polarizing beam splitter1110 into two beams having orthogonal polarization states designated byH and V in FIG. 9 b. It should be kept in mind that since the 1:Ksplitter 1107 is a linear device, it may instead be positioned eitherbefore the PMD compensator 1105 or the polarization beam splitter 1110.

The H-beam 1112 is directed to a variable delay 1115 controlled by asignal 1114 from the symbol synchronizer 1150. The variable delay 1115delays the H-beam so as to synchronize the H-beam to the V-beam 1111 sothat the H-beam and V-beam comprising the same symbol appear at thedemodulator at the same time. It should be understood that one mayinstead delay the V beam instead of the H beam.

The polarized H and V beams 1112, 1111 are first directed intodespreaders 1120 a, 1120 b, respectively, which leave the H and V beamsonly with phase information of the data used to modulate the code wordfor which the despreaders 1120 a, 1120 b are attuned. After despreading,the H beam is subjected to a first 1:2 splitter 1130 a, to formidentical beams H1 and H2, represented as signals 1131 a, 1132 a,respectively. After despreading, the V beam is subjected to a second 1:2splitter 1130 b to form identical beams V1 and V2, represented assignals 1131 b, 1132 b, respectively. Beams H2 and V2 and are directedinto symbol period delay circuits 1135 a, 1135 b, respectively, tothereby delay H2 and V2 by one symbol period. Beam H1 andone-symbol-period-delayed beam H2 are then input to a first opticalphase detector 1140 a, while beam V1 and one-symbol-period-delayed beamV2 are input to a second optical phase detector 1140 b. Thus, theoptical phase detectors 1140 a and 1140 b will output information aboutthe symbol-to-symbol phase difference, which is useful in differentialphase shift keying (DPSK).

The two optical phase detectors 1140 a, 1140 b output four digitalsignals designated as H_(i), H_(Q), V_(i), and V_(Q) that represent thein-phase and quadrature information of the H and V-beams. They alsogenerate the four analog signals H′_(i), H′_(Q), V′_(i), and V′_(Q),that correspond to the digital signals. The timing of the phasedetectors 1140 a, 1140 b is controlled by a timing signal 1242 from thesymbol synchronizer 1150. The four digital signals are input to the PMDcontroller/digital data modulator 1145, much as discussed above withrespect to the device 1045 of FIG. 10.

The structure of the Rx code despreaders 1120 a 1120 b, is substantiallythe same as the pulse spreader 330 shown in FIG. 4. The principledifference between an Rx code despreader 1120 a, 1120 b and the pulsespreader 330 is in the amplitude scaling and modulation applied to thepulse beam. For the k-th code despreader, the delay for the p-th chip,τ_(p)*, is the complement of the delay for the p-th chip in the k-thpulse spreader, τ_(p). Similarly, the Rx despreader phase shift,δ_(kp)*, is the conjugate phase shift applied by the pulse spreader 330and the despreader amplitude, a_(kp)*, is inverse of the amplitudescaling applied by the pulse spreader 330. The despreader delays andmodulations are given by the equations:τ_(p) *=T−τ _(p)  (2)δ_(kp)*=−δ_(kp)  (3)a _(kp)*=1/a _(kp)  (4)

Thus, within the K pairs of despreaders, each despreader 1120 a, 1120 bbelonging to k-th pair is attuned to the k-th code word. As discussedabove, the polarized H and V beams 1112, 1111 contain the entire codedivision multiplexed optical signal comprising all K orthogonal codewords modulated with data. When the H and V beams 1112, 1111 enter thek-th pair of despreaders, the output of that pair of despreaders is onlythe phase of the data modulated on the k-th code word, present at aknown location within the symbol. This is because the k-th pair ofdespreaders counters the code modulation of the k-th code word, leavingonly the phase of the data modulation in a known position (due to thetime-shifting of the time chips brought about by delay for the p-thchip, τ_(p)*), while the remaining K−1 code words, being orthogonal tothe k-th code word, more or less cancel out.

The despreaders 1120 a,1120 b are calibrated by calibration unit 1125which operates in a manner similar to that described with respect tospreader calibration units 335 or 335 a.

In the preferred embodiment using the coding scheme described byequation (1), the despreader delay and modulation for code k isidentical to the pulse spreader delay and modulation for code k. In apreferred embodiment, the upstream channel transmitter may be integratedwith the downstream channel receiver and vice versa. In such asituation, the imprinted code beams 322 generated by the pulse spreader330 for the upstream transmitter may be used as the K reference codebeams. For a generalized coding scheme encompassed by the presentinvention allowing both phase and amplitude modulation, however, thedespreader may not be identical to the pulse spreader.

FIG. 11 is a block diagram of the optical phase detectors 1030 a, 1030 bseen in the optical detection circuitry 1090 of FIG. 10 and the opticalphase detectors 1140 a, 1140 b seen in the channel receiver 260 a ofFIG. 9 b. The phase detector receives two optical beams 1705, 1701 andoutputs generates two electrical signals 1795, 1791 corresponding to thein-phase and quadrature components of the signal beam. One of the twobeams, designated as A in FIG. 7, is a signal beam. The second opticalbeam 1701, designated as B in FIG. 11, is a reference beam.

The signal beam 1705 is split into four beams identified by 1715 in FIG.11. In a preferred embodiment, the signal beam 1705 is split into fourbeams by a cascade of 1:2 splitters 1710. In another embodiment, thesignal beam 1705 may be directly split into four beams by a single 1:4splitter. Regardless of how they are formed, the four signal beams 1715are directed into four 2:1 combiners 1730 a, 1730 b, 1730 c, and 1730 d.

The reference beam 1701 is first subjected to a 1:2 splitter 1720 a toform identical beams R1 and R2. Beam R1 is then subjected to a second1:2 splitter 1720 b to form identical beams R3 and R4. Meanwhile, BeamR2 is first subjected to a 180° phase shifter 1724 before being spilt by1:2 splitter 1720 c to thereby form identical beams R5 and R6. Beam R3is input as signal 1721 to combiner 1730 a while beam R4 is firstsubjected to a first 90° phase shifter 1722 a before being input tocombiner 1730 b as signal 1723. Beam R5 is input as signal 1725 tocombiner 1730 c while beam R4 is first subjected to a second 90° phaseshifter 1722 b before being input to combiner 1730 d as signal 1727.

The resulting beams R3, R4, R5 and R6, represented as signals 1721,1723, 1725, and 1727 are phase-shifted by 90° increments. Beam R3/1721has zero phase shift and is combined with one of the four signal beams1715 in 2:1 combiner 1730 a to produce first combined beam 1731 that isthe sum of the signal and reference beam, designated as A+B. BeamR4/1723 has a 90° phase shift and is combined with one of the foursignal beams 1715 in second 2:1 combiner 1730 b to produce secondcombined beam 1733 that is designated as A+jB. Beam R5/1725 has a 180°phase shift and is combined with one of the four signal beams 1715 inthird 2:1 combiner 1730 c to produce third combined beam 1735 that isthe difference between the signal beam and reference beam and isdesignated as A−B. Finally, beam R6/1727 has a 270° phase shift and iscombined with one of the four signal beams 1715 in fourth combiner 1730d to produce fourth combined beam 1737 that is designated as A−jB. Thuscombiners 1730 a, 1730 b, 1730 and 1730 d are configured to combine acopy of signal A with a copy of signal B shifted by 0°, 90°, 180° and270°, respectively, to output first 1731, second 1733, third 1735 andfourth 1737 combined beams, respectively.

The first and third combined beams 1731, 1735 are input to a firstmatched detector 1740 a. The first matched detector 1740 a include lightsensors 1745 a, to thereby generate electrical signals that areproportional to the intensities of the first and third combined beams1731, 1735. The light sensors 1745 a are preferably photoelectricdetectors such as p-n, p-i-n, or Schottky-barrier photodiodes, and areselected to generate substantially identical electrical signals for thesame incident light beam. The electrical signals 1798 a generated by thefirst matched detector 1740 a are input to a first amplifier 1750 a. Theoutput signal 1752 of the first amplifier 1750 a is proportional to thein-phase difference between the signal beam 1705 and the reference beam1701.

The second and fourth combined beams 1733, 1737 are directed to a secondmatched detector 1740 b comprising matched light sensors 1745 b, tothereby generate electrical signals that are proportional to theintensities of the second and fourth beams 1733, 1737. The electricalsignals 1798 b generated by the second matched detector 1740 b are inputinto a second amplifier 1750 b. The output signal 1754 of the secondamplifier 1750 is proportional to the quadrature phase differencebetween the signal beam 1705 and the reference beam 1701. The splitters,phase shifters, combiners and detectors together comprise an optical 900hybrid detector 1799 that outputs the signals 1798 a, 1798 b.

The electrical signals 1798 a, 1798 b from the optical 900 hybriddetector 1799 are next input to a signal conditioning cascade circuitfor comprising an amplifier, a low pass filter, DC bias removal, sampleand hold circuitry and analog-to-digital converter. Signals 1798 a, 1798b are first amplified by amplifiers 1750 a, 1750 b, respectively. Theanalog output from the amplifiers 1750 a, 1750 b are input to low passfilters 1760 a, 1760 b, respectively. The low pass filters preferablyoperate at 40 GHz. The output of the lowpass filters is then subject todc bias adjustment 1770 a, 1770 b based on bias signals 1772, 1773,respectively, provided by a controller (not shown) which estimatessignal energy. The filtered and bias-adjusted analog signals I′ and Q′on lines 1775, 1777, respectively, are converted to digital form bysample and hold (“S&H”) units 1780 a, 1780 b, respectively, which aretimed by synchronization input 1781. The output of the sample and holdunits is sent on to analog-to-digital converters 1790 a, 1790 b to formthe in-phase I 1795 and quadrature Q 1791 signal which are subject tofurther processing, as is known to those skilled in the art.

The filtered and bias-adjusted analog signals I′ and Q′ on lines 1775,1777, respectively, are also sent on to the symbol synchronization units1040, 1150. Thus, the optical phase detector of FIG. 11 used inconjunction with the receiver is substantially similar to that shown inFIG. 7 for use on the transmitter side, with the exception of the analogtaps for the symbol synchronization units.

Symbol Synchronization

FIG. 12 is a block diagram of a preferred embodiment of the symbolsynchronization unit 1040/1150. Analog I′ and Q′ signals on lines 1775,1777 from both the H and V optical phase detectors, representing boththe in-phase and quadrature-phase components of the selected code beam,are each input into an early gate 1210 and a late gate 1215.

The early gate 1210 samples the signal a fraction of a symbol timebefore the S&H units 1780 a, 1780 b in the optical phase detector 1030a, 1030 b/1140 a, 1140 b perform their sampling. Conversely, the lategate 1215 samples the signal a fraction of a symbol time after the S&Hunits in the optical phase detector perform their sampling. In apreferred embodiment, the magnitude of the time offsets for the earlyand late gates are identical and equal to T/5 where T is the symbolperiod.

Each gate 1210, 1215 comprises a sample & hold unit followed by ananalog-to-digital converter, and is controlled by a clock signal fromthe clock distribution unit 1295. The signal from the clock distributionunit 1295 controls the sample & hold unit in each gate 1210, 1215 anddetermines the sampling instance of the input analog signals 1775, 1777by each gate 1210, 1215. The analog-to-digital converter converts theanalog sampled signal to digital format.

The in-phase and quadrature components from each of the gates 1210, 1215are squared 1220 and summed 1225 to determine the symbol energy for eachgate. The symbol energies are indicated in FIG. 12 as HE, HL, VE and VLwhere H and V designate the polarization states and E and L representthe early or late gate averages. The four symbol energies are input intoseparate delays 1230 and subtractors 1235. The delay 1230 delays thesignal representing the symbol energy by one symbol period beforesending the signal to the subtractor 1235. The subtractors 1235determine the differences between the preceding symbol energies and thecurrent symbol energies. The absolute value of the symbol-to-symbolenergy difference is taken in 1240. The difference between the earlygate and late gate energy difference is determined by subtractor 1245.

Averager 1250 averages the early/late gate symbol-to-symbol energydifference and inputs the output of the averager 1250 to a synchronizerloop filter 1270. The synchronizer loop filter 1270 controls a voltagecontrolled oscillator (VCO) 1280 that generates a timing signal 1242that is distributed by a clock distribution unit 1295 to the early andlate gates of the synchronizer and to the optical phase detectors.

Thus, with reference to FIG. 10, the symbol synchronizer 1040 receivesthe in-phase and quadrature components of the first and secondorthogonal polarization components of split beam signal 925, and outputsa timing signal 1242 to help synchronize the symbol boundaries of thedata-modulated codewords in the optical phase detectors 1030 a, 1030 b.

In the preferred implementation, the signal from the H and V channelsare averaged to control a single VCO. In another implementation, onlyone channel is used to control the VCO. In yet another implementation,each of the channels may control a different VCO.

Subtractor 1255 determines the difference between the symbol-to-symbolenergy differences from the H and V optical phase detectors. Thedifference between the H and V optical phase detectors is input into thedelay loop filter 1260. The delay loop filter 1260 generates a controlsignal 1014/1114 to the H-beam variable delay 1015/1115 based on thesymbol-to-symbol energy difference between the H and V optical phasedetectors.

In the above implementation of the symbol synchronizer 1040, 1150, it isnoted that the various computations are preferably performed in softwareby a DSP, ASIC or another, preferably programmable, processor.

FIGS. 13 a and 13 b illustrate the advantage of using thesymbol-to-symbol energy difference, instead of using just the symbolenergy. FIG. 13( a) is a graph of the average symbol energy 1310 as afunction of the normalized symbol time. A symbol time of zerocorresponds to the beginning of a symbol. An energy envelope 1315approximately indicates the variation that can be expected in the symbolenergy measurement as a function of symbol time. The energy envelopeexists because the signal still contains the signals from the othercombined codes of the channel. The average symbol energy 1310 exhibits aslight maximum 1320 when the symbol time is synchronized to the symbol.The slight maximum makes it very difficult to locate the maximum in thesymbol energy curve.

FIG. 13( b) is a plot of the average symbol-to-symbol energy difference1350 as a function of the symbol time. An energy envelope 1360approximately indicates the variation that can be expected in thesymbol-to-symbol energy difference measurement as a function of symboltime. The average symbol-to-symbol energy difference 1350 exhibits arelatively deep minimum 1370 when the symbol time is synchronized to thesymbol. The relatively deep minimum allows the loop filter to locate andkeep the symbol time at the minimum 1370.

Polarization Compensation

FIG. 14 a is a block diagram of a communication system 1400 that is ableto adjust the SOP of the transmitted signal such that the transmittedsignal is launched along the axes of the fiber 1474 associated with theminimum spreading of the received signal, while also compensating forthe received PMD. In theory, the fiber exhibits two orthogonaleigenstates that are independent of frequency and fiber length. When thesignal is launched along these eigenstates (fast and slow axes) of thefiber, PMD is limited to the differential group velocities of the two,and coupling of the signal's components traveling along each of the axesis zero. In practice, there are no such eigenstates for the long fiber,but for each optical frequency there exist two orthogonal inputpolarization states for which the output polarization states exhibit theleast frequency dependence. Such states are called the principle stateof polarization (PSP) of the fiber; the signal launched along the PSPexhibits the least coupling and therefore, the minimal temporalspreading. Due to environmental variations along the fiber with time,the PSP also varies with time and the variations should be tracked andcompensated for.

The communication system 1400 includes a transmitter 1470 that directsthe multiplexed and modulated optical signal to a transmitterpolarization compensator 1472 that adjusts the transmitter SOP (Tx SOP)before launching the optical signal through the optical fiber 1474. Atthe receiver, a receiver polarization compensator 1476 adjusts thereceiver SOP (Rx SOP) to optimize the selected metric. The receiver 1478includes a PMD controller control both the receiver polarizationcompensator 1476 via signal line 1477 and the transmitter polarizationcompensator 1472 via a supervisory channel 1473 (represented by line 295in FIGS. 2 a and 2 b and lines 1056, 1156 in FIGS. 9 a, and 9 b,respectively). The compensators 1472 and 1476 are discrete devices suchas described above. The PMD controller may be implemented in hardwarebut in a preferred embodiment, the controller is implemented as acomputer program executing on a processor such as a DSP.

FIG. 14 b presents a flow diagram 1405 describing the operation of thePMD controller Rx module for controlling the Rx SOP. The PMD controllerdithers the receiver PMD compensator and executes a search to determinethe compensation required to optimize some metric. In a preferredembodiment, the metric is the symbol signal-to-noise-ratio (SNR), andthis is to be maximized. It should be noted, however, that one mayinstead use the error vector magnitude (EVM) as the metric, and minimizethis. Other metrics may also be employed.

In step 1410, the dither step parameters, which include dither size anddither direction are established. The size and direction are establishedwith respect to movement on the surface of a Poincare′ sphererepresenting the receiver's SOP. The step directions are with respect torotations, Δθ and Δε, about the H-V and P-Q axes, respectively. Eachdither step comprises a pair of rotations. Each rotation can take threepossible values (±Δθ and 0 for the H-V rotation and ±Δε and 0 for theP-Q rotation) resulting in a total of nine dither steps. Thus,establishing the dither step parameters entails selecting one from amongnine candidate steps.

In step 1420, the receiver's PMD compensator is dithered by one ofcandidate steps by applying the proper set of rotations. In step 1430,the metric to be optimized is estimated by the controller and storedalong with the state of the receiver PMD compensator 1476, the statebeing reflective of the candidate step that had just been tried.Techniques for estimating a metric such as the SNR are well-known andstandard algorithms for this can be executed by the processor. Thesignals used for this preferably are known pilot signals sent by thetransmitter, and the receiver knows what to expect.

In step 1440, a check is made to determine whether all candidates havebeen tried. If not, control returns to step 1410 to try the nextcandidate step. If, on the other hand, all candidate steps have beentried, control flows to step 1450. In step 1450, the candidate step/PMDcompensator state corresponding to the optimized metric (e.g., highestSNR) is determined and in step 1460, the PMD compensator is adjusted tothe optimized state/location on the Poincare′ sphere. For this, the PMDcontroller commands the receiver PMD compensator 1476 through signalline 1477 to execute the optimal adjustment in 1460. Preferably, theabove process is repeated until some terminal condition is met. Theterminal condition can be convergence, i.e., there are no more changesin state, or upon completion of a predetermined number of ditheringloops. The desirable result from this process is to restore the signalpolarization components by re-aligning the RX SOPs with the TX SOPs.

While the above description calls for an exhaustive search through allpossible candidate steps, it should be kept in mind that fewer than allpossible steps can be tried. In a preferred embodiment, the ditheradjustment is determined by a steepest gradient method. This allows oneto reduce processing time (such as by choosing larger step sizes) inexchange for a slightly elevated risk that one misses the global optimumon the Poincare′ sphere, when searching for an optimum metric.

After determining the optimal adjustment, The PMD controller Rx moduledescribed above with respect to FIG. 14 b can find the optimum state forthe receiver PMD compensator 1476 to compensate for the SOPtransformation taking place in the fiber at the receiver end of thesystem. This will not, however, compensate for spreading of the signalcaused by the frequency dependence of the transmitted signal's SOP if itdoes not coincide with the PSP axes of the fiber. For this, one mustalso compensate the transmitted signal, before it is transmitted.

FIG. 14 c presents a flow diagram 1480 describing the operation of thePMD controller Rx/Tx module to control both the receiver PMD compensator1476 and the transmitter SOP compensator 1472, in a system such as thatseen in FIG. 14 a.

In step 1482, the PMD controller adjusts the Rx compensator to re-alignthe Rx and Tx SOPs by dithering only the receiver, as discussed abovewith respect to flow diagram 1405 in FIG. 14 b. This means that theincoming signal is being properly compensated at the receiver.

Beginning with step 1484, the controller next searches for the PSP ofthe fiber by examining the ensemble of the candidate SOPs on thePoincare′ sphere for the transmitter. In this regard, it is noted thatthe Tx SOP candidates are assumed to be uniformly distributed on thetransmitter's Poincare′ sphere with certain resolution—i.e., eachcandidate is “assigned” some portion of the sphere; the larger thenumber of candidates, the smaller the portion size.

Thus, in step 1484, a candidate Tx SOP is selected, and the Txcompensator is adjusted accordingly.

In step 1486, the receiver SOPs are adjusted to conform the newcandidate Tx SOP so as to preserve the TX/RX SOP alignment. This is doneby adjusting the receiver PMD compensator in a manner complementary tothe adjustment made to the transmitter compensator. In effect then, arotation of the Tx compensator 1472 to a new candidate SOP is followedby a “counter-rotation” of the Rx compensator 1476. To ensure synchronybetween the transmitter and the receiver, the Tx SOPs preferably areadjusted on some predetermined basis that is known to the receiver so asto allow the Rx SOPs to be changed each time a new Tx candidate SOP istried.

In step 1488, the relevant metric is calculated and stored, along withthe states of the Tx SOPs and the Rx SOPs. The optimal TX SOP (i.e., thePSP) are chosen based on the same criteria as used by the RX compensator(e.g., maximum SNR).

In step 1490, a check is made to determine whether all the candidate TxSOPs have been evaluated. If it is determined in step 1490 thatadditional candidate Tx SOPs remain, these are tried. If, on the otherhand, it is determined in step 1490 that there are no further candidateTx SOPs, controls flows to step 1492.

In step 1492, the transmitter and receiver compensators 1472, 1476,respectively, are adjusted based on the states corresponding to theoptimum metric. Thus, the final TX/RX SOP adjustment takes place toalign the TX SOP with the PSP axes.

The above-described procedure is repeated to track the polarizationchanges occuring in the fiber. Preferably, this is done on a continuousbasis. The candidate PSP list can be limited to the subset of theoriginal list situated around the current PSP axes during the tracking.This limitation is based on the assumption that the small variations inthe fiber polarization transformation matrix will lead to smallvariations of the PSP orientation.

In the above-described operation of the PMD controller, the Receiver andTransmitter compensators are preferably implemented using theaforementioned Acrobat™ Polarization Control Module (PCM). It should bekept in mind, however, that other adjustable polarization controldevices may be used, as well.

Transmitter Having Multiple Coding Stages

FIG. 15 is a block diagram 1500 of an alternative channel transmitterconfigured to create a code division multiplexed signal in accordancewith the present invention. The channel transmitter 1500 creates thecode words in two stages: a pre-coding stage 1502 and a final codingstage 1504. In the following description, K represents the total numberof code words that are created, P represents the number of time chips ina code, and L represents the number of code words created from eachpre-coding stage. In a preferred embodiment described below, K=16, P=16and L=4, and the symbol period T=400 ps.

The channel transmitter 1500 includes a pulsed light source 1510 whichgenerates an optical pulse stream 1515. The optical pulse stream 1515preferably comprises pulses separated by T*UK=100 ps, given thepreferred parameters. Thus, an optical pulse stream from source 1510having a symbol length T=400 ps has a total of 4 pulses, each pulsebeing found in a single 100 ps window. This contrasts with the pulsespacing of 400 ps provided by the pulsed light source 310 in theembodiment of FIG. 3. Accordingly, the pulsed light source 1510 outputspulse 4 times as fast as pulsed light source 310.

The optical pulse stream 1515 is directed into a 1:K/L splitter 1520 toproduce K/L identical optical pulse streams 1525. Each identical opticalpulse stream 1525 is then subject to a separate dynamic code modulator(DCM) 1530 a, 1530 b, 1530 c, 1530 d to create a corresponding sub-codebeam 1535 a, 1535 b, 1535 c, 1535 d. Each DCM imparts a first phaseshift to each of the pulses input thereto, based on pre-determinedcontrol signals from a DCM controller 1532. The resulting sub-code beams1535 a, 1535 b, 1535 c, 1535 d preferably are orthogonal to one another.For clarity, further processing applied to only one sub-code beam 1535 ais discussed, it being understood that the remaining sub-code beams 1535b, 1535 c and 1535 d undergo the same processing.

The sub-code beam 1535 a is directed into a 1:L splitter to form fouridentical sub-code beams 1545 a, 1545 b, 1545 c, 1545 d having the sameset of phase shifts. In general, L represents the number of code wordscreated by a sub-code beam. Thus, each of the L=4 identical sub-codebeams is then directed into a corresponding pulse spreader 1550 a, 1550b, 1550 c, 1550 d. Pulse spreaders 1550 a, 1550 b, 1550 c, 1550 d areconfigured work substantially the same as pulse spreader 330 seen inFIG. 4, except that the spreader splitter performs a 1:PLK split (PL/Kbeing an integer) and the spreader combiner forms a PUK:1 combination.Thus, for example, pulse spreader 1550 a includes a 1 :PL/K splitter,delay circuitry for PL/K delays, chip modulation circuitry, and a PL/K:1combiner. A spreader calibration unit 1585 controls the spreaders, in amanner analogous to spreader calibration units 335, 335 a, describedabove. And for P=16, K=16, L=4; PL/K=4, as seen in FIG. 15.

A sub-code beam 1545 a comprising a 400 ps-long symbol of four pulsesspaced apart from one another by 100 ps enters the pulse spreader 1550a. Within pulse spreader 1550 a, the first of the four pulses arrives inthe first 100 ps window and is subject to a 1:4 split to create 4identical copies. Delays of 0 ps, 25 ps, 50 ps and 75 ps are applied toeach of the 4 split pulses, a chip modulation circuit then applies asecond phase shift (or applies both a second amplitude and phase) toeach of the four spilt pulses, and the code-modulated four split pulsesare recombined in a 4:1 combiner. This results in four code-modulatedpulses within the first 100 ps window, each pulse having been modulatedtwice: once by the dynamic code modulator 1530 a in the first stage1502, and a second time by a chip modulating circuit associated with thepulse spreader 1550 a in the second stage 1504. The second, third andfourth pulses belonging to sub-code beam 1545 a arrive in the second,third and fourth 100 ps windows of the symbol, respectively. They aresimilarly split, delayed, subject to a second phase shift by the samepulse spreader 1550 a, and then recombined.

Thus, the output of the pulse spreader 1550 a is thus a single code word1555 a having a total of P=16 pulses. Similarly, the output of the otherpulse spreaders 1550 b, 1550 c, 1550 d are different code words 1555 b,1555 c, 1555 d, respectively, each of the four code words originatingfrom the sub-code word 1535 a being orthogonal to the other three, andalso to the other 12 code words originating from the other threesub-code words 1535 b, 15335 c, 1535 d.

The ensemble of code words next enters a data modulation stagecomprising data modulators 1560 a, 1560 b, 1560 c, 1560 d. Code word1555 a enters the data modulator 1560 a where the data from a datastream 1570 under the control of a data controller (not shown) isimprinted thereon to thereby form a data-modulated code word 1565 a.This process is repeated for code words 1555 b, 1555 c, 1555 d with datamodulators 1560 b, 1560 c, 1560 d, respectively, to form data-modulatedcode words 1565 b, 1565 c, 1565 d, respectively. These fourdata-modulated code words, along with the other 12 code wordsoriginating from the other three sub-code words 1535 b, 15335 c, 1535 d(for a total of K=16 code words), are then input to a 1:K combiner 1580to create the data beam 1575 which is directed into a DWDM multiplexer(not shown). The data beam is thus a division multiplexed optical signalcomprising orthogonal code words that have been data-modulated, the codewords preferably having H and V polarizations due to a polarization beamcombiner within the data modulators.

The transmitter 1500 is provided with one or more spreader calibrationunits 1585, which behave in a manner similar to those discussed above.Thus, the spreader calibration units 1585 receive inputs from areference light source 1518 and also from the outputs of the pulsespreaders 1555 a, 1555 b, 1555 c, 1555 d on lines 1556. The spreadercalibration units 1585 also output signals to these pulse spreaders, toensure that each of the pulse spreaders has proper modulation.

The invention described and claimed herein is not to be limited in scopeby the preferred embodiments herein disclosed, since these embodimentsare intended as illustrations of several aspects of the invention. Anyequivalent embodiments are intended to be within the scope of thisinvention. Indeed, various modifications of the invention in addition tothose shown and described herein will become apparent to those skilledin the art from the foregoing description. Such modifications are alsointended to fall within the scope of the appended claims.

1. A code division multiplexed optical communication system comprising:at least one transmitter configured to receive a first data stream andtransmit a code division multiplexed optical signal comprising a numberK code words modulated with first data from said first data stream(whereK is an integer greater than 1); wherein the transmitter comprises: apulsed light source: a transmitter splitter having a splitter input anda plurality of splitter outputs, the transmitter splitter having thepulsed light source input thereto and outputting at least a number Kidentical code beams; K code modulators, each code modulator configuredto receive one of the at least K identical code beams and output acorresponding data-modulated code word; wherein each of the codemodulators comprises: a pulse spreader configured to receive one of theK identical code beams, said one of the K identical code beamscomprising a single pulse within a predetermined time window, the pulsespreader being further configured to output an imprinted code beamhaving a number P modulated pulses within that time window; and a datamodulator configured to receive said imprinted code beam from the pulsespreader and modulate said imprinted code beam with data from said firstdata stream to thereby form a data-modulated code word; wherein thepulse spreader comprises: a 1:P splitter configured to split one of saidK code beams input thereto into P identical code beams (where P is aninteger greater than 1), each code beam having a single pulse; P chipmodulators, each chip modulator configured to receive a single pulse andoutput a delayed modulated pulse, the p-th chip modulator comprising: adelay circuit configured to delay said single pulse by (p-1)*C, whereC=T/P is a chip period, T being a symbol period (T is an integer greaterthan 1) and p representing an index; and a chip modulation circuitconfigured to code-modulate the delayed single pulse by a p-th codevalue belonging to an orthogonal code of length P; and a P:1 combinerconfigured to combine code-modulated outputs from the P chip modulationcircuits into the imprinted code beam which comprises P pulses withinsaid symbol period T; and a transmitter combiner configured to combinethe K data-modulated code words into a code division multiplexed opticalsignal; wherein the K data-modulated code words are orthogonal to oneanother; and at least one receiver optically connected to the firsttransmitter and configured to receive said code division multiplexedoptical signal, detect and demodulate said K code words within the codedivision multiplexed optical signal, and output said first data, whereinthe K code words are orthogonal to one another.
 2. The opticalcommunication system of claim 1, wherein the transmitter furthercomprises a spreader calibration unit which receives the imprinted codebeam from the pulse spreader and a reference light source as inputs, andoutputs a spreader control signal which is sent to the pulse spreader.3. The optical communication system of claim 1 wherein the datamodulator comprises: splitter circuitry configured to split theimprinted mode beam into identical first (H1), second (H2), third (V1)and fourth (V2) component beams; a first phase shifter configured toimpart a 90.degree. phase shift the second component beam (H2); a secondphase shifter configured to impart a 90.degree. phase shift to thefourth component beam (H4); a first modulator configured to modulate thefirst component beam (H1) with first data; a second modulator configuredto modulate the phase-shifted second component beam (H2) with seconddata; a third modulator configured to modulate the third component beam(V1) with third data; a fourth modulator configured to modulate thephase-shifted fourth component beam (V2) with fourth data; a firstcombiner (840 a) configured to combine the data-modulated firstcomponent beam (H1) with the data-modulated and phase-shifted secondcomponent beam (H2), and output a first data-modulated beam (H′); asecond combiner (840 b) configured to combine the data-modulated thirdcomponent beam (V1) with the data-modulated and phase-shifted fourthcomponent beam (V2), and output a second data-modulated beam (V′); and apolarization beam combiner (850) configured to combine the first andsecond data-modulated beams and output a data beam (342) having twoorthogonal polarizations.
 4. The optical communication system of claim 1wherein the receiver comprises: a receiver splitter having a splitterinput and a plurality of splitter outputs, the splitter input receivingthe code division multiplexed optical signal comprising K data-modulatedcode words, and outputting at least K identical received code divisionmultiplexed optical signals each comprising K data-modulated code words;and at least K code receivers, each code receiver having one of saididentical received code division multiplexed optical signals inputthereto, each code receiver having associated therewith: a receiverpulse spreader configured to create a reference imprinted code beamcorresponding to one of the K code words in said received code divisionmultiplexed optical signal; and a receiver unit having an informationsignal and a reference signal input thereto, wherein the informationsignal is said one of said identical received code division multiplexedoptical signals and the reference signal is the imprinted reference codebeam, the receiver unit configured to detect and demodulate said one ofthe K data-modulated code words to which the reference imprinted codebeam corresponds.
 5. The optical communication system of claim 4,wherein: the receiver unit comprises an optical detection circuit thatreceives the information signal and the reference signal, the opticaldetection circuit configured to output in-phase and quadraturecomponents of a first and a second orthogonal polarization component ofthe information signal.
 6. The optical communication system of claim 5,wherein the receiver unit further comprises a polarization modedispersion controller configured to: receive digitized in-phase andquadrature components of the first and second orthogonal polarizationcomponents of the information signal, and output at least onepolarization control signal in response thereto.
 7. The opticalcommunication system of claim 6, wherein the at least one polarizationcontrol signal is input to a polarization compensator associated withthe receiver.
 8. The optical communication system of claim 5, whereinthe optical detection circuit comprises: a polarization beam splitterconfigured to receive and split the information signal into first andsecond orthogonal polarization components; a first optical phasedetector configured to receive the first orthogonal polarizationcomponent and the reference signal as inputs, and output in-phase andquadrature components of the first orthogonal polarization component; asecond optical phase detector configured to receive the secondorthogonal polarization component and the reference signal as inputs,and output in-phase and quadrature components of the second orthogonalpolarization component.
 9. The optical communication system of claim 8,wherein the optical detection circuit further comprises: a symbolsynchronizer circuit receiving said in-phase and quadrature componentsof the first and second orthogonal polarization components of theinformation signal, and outputting at least one timing signal tosynchronize symbol boundaries of said data-modulated codewords in saidoptical phase detectors. a variable delay configured to synchronize thefirst and second orthogonal polarization components, the variable delayreceiving a delay signal from the symbol synchronizer circuit.
 10. Theoptical communication system of claim 8, wherein the first and secondoptical phase detectors each comprise: an optical hybrid detector havingfirst and second signal inputs and first and second signal outputs, thefirst signal output being proportional an in-phase difference betweenthe first and second signal inputs, and the second signal output beingproportional to a quadrature difference between the first and secondsignal inputs; and a first signal conditioning cascade circuitcomprising a first amplifier, a first low pass filter, a first DC biasremover, a first sample and hold, and a first analog-to-digitalconverter, all arranged to process the first signal output from theoptical hybrid detector to thereby form a digitized in-phase componentsignal; and a second signal conditioning cascade circuit comprising asecond amplifier, a second low pass filter, a second DC bias remover, asecond sample and hold, and a second analog-to-digital converter, allarranged to process the second signal output from the optical hybriddetector to thereby form a digitized quadrature component signal. 11.The optical communication system of claim 10, wherein the optical hybriddetector having first and second signal inputs comprises: opticalcircuitry configured to split and phase shift the second signal input toform a first signal (R3) having 0.degree. phase shift, a second signal(R4) having a 90.degree. phase shift, a third signal (R5) having a180.degree. phase shift and a fourth signal (R6) having a 270.degree.phase shift; first, second, third and fourth combiners, configured tocombine a copy of the first signal input with a copy of the secondsignal input shifted by 0.degree., 90.degree., 180.degree. and270.degree., respectively, to output first second, third and fourthcombined beams, respectively; a first matched detector configured toreceive said first and third combined beams and output a first outputsignal that is proportional to an in-phase difference between the firstand second signal inputs; and a second matched detector configured toreceive said second and fourth combined beams and output a second outputsignal that is proportional to quadrature difference between the firstand second signal inputs.
 12. A pulse spreader configured to accept acode beam comprising a single pulse within a symbol period of length T,and output an imprinted code beam comprising a number P modulated pulseshaving a length C=T/P chip periods, the pulse spreader comprising: a 1:Psplitter configured to split the code beam input thereto into Pidentical code beams, each code beam having a single pulse; P chipmodulators, each chip modulator configured to receive a single pulse andoutput a delayed modulated pulse, the p-th chip modulator comprising: adelay circuit configured to delay said single pulse by (p−1)*C, whereC=T/P is a chip period, p representing an index; and a chip modulationcircuit configured to code-modulate the delayed single pulse by a p-thcode value belonging to an orthogonal code of length P; and a P:1combiner configured to combine code-modulated outputs from the P chipmodulation circuits into the imprinted code word comprising P pulseswithin said symbol period T.
 13. An optical signal data modulator formodulating an input signal, the optical signal data modulatorcomprising: optical splitter circuitry configured to split the inputsignal into identical first (H1), second (H2), third (V1) and fourth(V2) component beams; a first phase shifter configured to impart a90.degree. phase shift to the second component beam (H2); a second phaseshifter configured to impart a 90.degree. phase shift to the fourthcomponent beam (H4); a first modulator configured to modulate the firstcomponent beam (H1) with first data; a second modulator configured tomodulate the phase-shifted second component beam (H2) with second data;a third modulator configured to modulate the third component beam (V1)with third data; a fourth modulator configured to modulate thephase-shifted fourth component beam (V2) with fourth data; a firstcombiner (840 a) configured to combine the data-modulated firstcomponent beam (H1) with the data-modulated and phase-shifted secondcomponent beam (H2), and output a first data-modulated beam (H′): asecond combiner (840 b) configured to combine the data-modulated thirdcomponent beam (V1) with the data-modulated and phase-shifted fourthcomponent beam (V2), and output a second data-modulated beam (V′); and apolarization beam combiner (850) configured to combine the first andsecond data-modulated beams and output a data beam (342) having twoorthogonal polarizations.
 14. The optical signal data modulator of claim13, wherein the first and second phase shifters and the first, second,third and fourth modulators receive control signals from a powerbalancer.
 15. A code division multiplexed optical signal receivercomprising: a receiver splitter having a splitter input and a pluralityof splitter outputs, the splifter input receiving the code divisionmultiplexed optical signal comprising K code words modulated with data,and outputting at least K identical received code division multiplexedoptical signals each comprising K data-modulated code words; and atleast K code receivers, each code receiver having one of said identicalreceived code division multiplexed optical signals input thereto, eachcode receiver having associated therewith: a receiver pulse spreaderconfigured to create a reference imprinted code beam corresponding toone of the K code words in said received code division multiplexedoptical signal; and a receiver unit having an information signal and areference signal input thereto, wherein the information signal is saidone of said identical received code division multiplexed optical signalsand the reference signal is the imprinted reference code beam, thereceiver unit configured to detect and demodulate said one of the Kdata-modulated code words to which the reference imprinted code beamcorresponds.
 16. An optical phase detector in optical communicationssystem comprising: an optical hybrid detector having first and secondsignal inputs and first and second signal outputs, the first signaloutput being proportional an in-phase difference between the first andsecond signal inputs, and the second signal output being proportional toa quadrature difference between the first and second signal inputs; anda first signal conditioning cascade circuit comprising a firstamplifier, a first low pass filter, a first DC bias remover, a firstsample and hold, and a first analog-to-digital converter, all arrangedto process the first signal output from the optical hybrid detector tothereby form a digitized in-phase component signal; and a second signalconditioning cascade circuit comprising a second amplifier, a second lowpass filter, a second DC bias remover, a second sample and hold, and asecond analog-to-digital converter, all arranged to process the secondsignal output from the optical hybrid detector to thereby form adigitized quadrature component signal.
 17. The optical phase detector ofclaim 16, wherein the optical hybrid detector having first and secondsignal inputs comprises: optical circuitry configured to split and phaseshift the second signal input to form a first signal (R3) having0.degree. phase shift, a second signal (R4) having a 90.degree. phaseshift, a third signal (R5) having a 180.degree. phase shift and a fourthsignal (R6) having a 270.degree. phase shift first, second, third andfourth combiners, configured to combine a copy of the first signal inputwith a copy of me second signal input shifted by 0.degree., 90.degree,180.degree and 270.degree, respectively, to output first, second, thirdand fourth combined beams, respectively; a first matched detectorconfigured to receive said first and third combined beams and output afirst output signal that is proportional to an in-phase differencebetween the first and second signal inputs; and a second matcheddetector configured to receive said second and fourth combined beams andoutput a second output signal that is proportional to quadraturedifference between the first and second signal inputs.
 18. An opticalhybrid detector of claim 16 having first and second signal inputs andcomprising: optical circuitry configured to split and phase shift thesecond signal input to form a first signal (R3) having 0.degree. phaseshift, a second signal (R4) having a 90.degree. phase shift, a thirdsignal (R5) having a 180.degree. phase shift and a fourth signal (R6)having a 270.degree. phase shift; first, second, third and fourthcombiners, configured to combine a copy of the first signal input with acopy of the second signal input shifted by 0.degree., 90.degree.,180.degree. and 270.degree., respectively, to output first, second,third and fourth combined beams, respectively; a first matched detectorconfigured to receive said first and third combined beams and output afirst output signal that is proportional to an in-phase differencebetween the first and second signal inputs; and a second matcheddetector configured to receive said second and fourth combined beams andoutput a second output signal that is proportional to quadraturedifference between the first and second signal inputs.